DCM, BCM & CCM Flyback Operation | Flyback Design Guide

A flyback converter may operate in discontinuous conduction mode, boundary conduction mode, or continuous conduction mode. These modes are defined by the magnetizing-current waveform and by whether the transformer fully demagnetizes before the next switching cycle begins. The selected mode affects peak and RMS current, transformer energy storage, core bias, winding loss, semiconductor stress, control dynamics, EMI, and the practical size of the magnetic component.

By the end of this chapter, you should be able to identify DCM, BCM, and CCM from measured waveforms; calculate the current and timing conditions that define each mode; understand how operating mode changes transformer design; and select a mode or multimode strategy that fits the converter power level, control method, efficiency target, and operating range.

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DESIGN ASSUMPTION — IDEALIZED MODE COMPARISON

The first comparisons use an idealized flyback power stage with fixed input voltage, fixed output voltage, ideal coupling, negligible winding resistance, negligible leakage inductance, and linear current ramps. The current variable used to identify operating mode is the magnetizing current referred to the primary side.

Real controllers may vary switching frequency, skip cycles, enter burst mode, select valley switching points, or transition automatically between DCM and CCM. Those behaviors are addressed later in the chapter. DCM, BCM, and CCM describe the magnetic-current condition; they do not by themselves specify the complete control strategy.

1. What Conduction Mode Means in a Flyback Converter

Conduction mode describes the magnetizing-current state at the boundary between switching cycles. In an ideal flyback converter, the primary winding carries current during the MOSFET ON interval and the secondary winding carries the corresponding magnetic current during the OFF interval. A continuous primary-referred magnetizing-current waveform can be used to describe the magnetic state across both intervals, even though physical primary current is zero after the MOSFET turns OFF.

The three operating modes are distinguished by the minimum magnetizing current and by the presence or absence of a zero-current interval:

Operating Mode	Magnetizing-Current Condition	Cycle Boundary
DCM	Current reaches zero before the next cycle	A finite zero-current interval remains.
BCM / CrM / TM	Current reaches zero at approximately the next turn-on instant	The zero-current interval approaches zero.
CCM	Current remains above zero at the next cycle	Residual magnetizing current carries into the following cycle.

Let D represent the primary ON-time fraction, D_S the secondary conduction fraction, and D_Z the zero-current fraction. For one idealized switching period:

D+D_S+D_Z=1

In CCM, D_Z is zero and the secondary current remains nonzero at the end of the OFF interval. In BCM, D_Z approaches zero while the current itself reaches zero at the boundary. In DCM, D_Z is positive.

SolidMag Engineering Insight

Operating Mode Is a Property of the Current Waveform

A controller may be marketed as fixed-frequency, quasi-resonant, valley-switching, or multimode, but the transformer still operates in DCM, BCM, or CCM at each individual operating point according to its magnetizing-current waveform.

The magnetic design should therefore be checked over the complete line and load range rather than assigning one conduction mode to the product without verifying the actual current at every important operating condition.

2. The Magnetizing-Current Waveform Defines the Mode

During the primary ON interval, the current change is determined by the primary winding voltage, magnetizing inductance, duty cycle, and switching frequency:

\Delta I_P=\frac{V_{P,ON}D}{L_m f_s}

For a linear current ramp, the peak and minimum values are related by:

I_{MAX}=I_{MIN}+\Delta I_P

The secondary current is the same magnetic-current state scaled by the turns ratio. With n = N_P/N_S, the idealized secondary current is approximately n times the primary-referred magnetizing current during the OFF interval.

A mode decision should be based on the current immediately before the next primary turn-on:

DCM: I_MIN = 0 and a finite interval exists before the next cycle.
BCM: I_MIN approaches zero at the next switching boundary.
CCM: I_MIN > 0 at the next switching boundary.

Comparison of DCM, BCM, and CCM flyback operation showing aligned primary current waveforms and how discontinuous, boundary, and continuous conduction modes differ over the switching cycle.

3. Discontinuous Conduction Mode — DCM

In discontinuous conduction mode, all incremental magnetizing energy is delivered before the next switching cycle begins. The primary-referred magnetizing current starts near zero, rises to a peak during the ON interval, decreases to zero during secondary conduction, and then remains at zero for the rest of the cycle.

I_{MIN}=0

D_Z>0

DCM Peak Current and Energy

Because the current starts near zero, the peak primary current is approximately equal to the ON-time current change:

I_{P,PK}\approx\Delta I_P=\frac{V_{P,ON}D}{L_m f_s}

The peak stored magnetizing energy is:

E_{PK}=\frac{1}{2}L_m I_{P,PK}^{2}

If that energy is processed once per cycle, the idealized input power is:

P_{IN}\approx\frac{1}{2}L_m I_{P,PK}^{2}f_s

DCM RMS Currents

For a linear primary-current ramp from zero to I_P,PK during duty cycle D:

I_{P,RMS}=I_{P,PK}\sqrt{\frac{D}{3}}

If the secondary current begins at I_S,PK and decays linearly to zero over D_S:

I_{S,RMS}=I_{S,PK}\sqrt{\frac{D_S}{3}}

Typical DCM Advantages and Tradeoffs

DCM Tendency	Engineering Consequence
Complete demagnetization each cycle	Secondary rectifier current reaches zero before the next primary turn-on.
Finite zero-current interval	Can support valley switching and reduced turn-on loss in suitable controllers.
Higher peak current for a given power	Increases MOSFET, current-sense, winding, rectifier, and clamp stress.
Higher pulsed RMS current	Can increase copper loss and capacitor ripple current.
Zero-current secondary turn-off	Reduces diode reverse-recovery stress relative to deep CCM.
Simpler small-signal behavior	Avoids the CCM right-half-plane-zero limitation in the usual DCM control model.

ENGINEERING CAUTION — DCM DOES NOT AUTOMATICALLY MEAN LOW EMI

DCM can reduce reverse-recovery stress, but its higher peak current and fast current transitions can increase ringing, clamp energy, common-mode current, and radiated or conducted emissions.

EMI depends on the complete switching loop, transformer parasitics, clamp behavior, layout, edge rate, shielding, and filtering—not on conduction mode alone.

4. Boundary Conduction Mode — BCM, CrM, or Transition Mode

Boundary conduction mode is the exact transition between discontinuous and continuous operation. It is also called critical conduction mode, CrM, or transition mode. The magnetizing current reaches zero at approximately the instant the next primary switching cycle begins.

I_{MIN}\approx0

D_Z\approx0

D+D_S\approx1

At the boundary, the transformer uses essentially the entire cycle for energy storage and delivery without carrying residual current into the next cycle. Many BCM or quasi-resonant controllers vary switching frequency so the next turn-on occurs after demagnetization, often at a selected drain-voltage valley.

Boundary Magnetizing Inductance at a Defined Operating Point

For an idealized boundary operating point at a defined frequency, the DCM energy equation and the current-ramp equation can be combined to estimate the magnetizing inductance that places the converter at the boundary:

L_{m,BCM}=\frac{V_{P,ON}^{2}D^{2}}{2P_{IN}f_s}

This is an operating-point relationship, not a universal transformer value. If line voltage, load, duty cycle, or switching frequency changes, the same transformer can move away from the boundary into DCM or CCM.

Typical BCM Advantages and Tradeoffs

No intentional zero-current dwell interval at the boundary operating point.
Secondary current reaches zero naturally, which can reduce rectifier recovery stress.
Variable-frequency operation can support valley switching and reduce MOSFET turn-on loss.
The switching-frequency range can become wide across line and load, complicating EMI filters, audible-noise control, magnetic-loss prediction, and timing limits.
Peak current remains higher than in many CCM designs delivering the same power.

SolidMag Engineering Insight

BCM Is a Boundary Condition, Not a Wide Operating Region

Only one combination of line, load, frequency, duty cycle, and magnetizing inductance is exactly at the boundary. A real converter normally crosses through BCM while moving between DCM and CCM or actively adjusts its switching frequency to remain near the boundary.

When a design is described as BCM, verify whether the controller actually regulates at the boundary across the intended range or merely passes through it at one operating point.

5. Continuous Conduction Mode — CCM

In continuous conduction mode, the magnetizing current does not return to zero before the next primary ON interval. The next cycle begins with a nonzero minimum current, so the transformer retains stored magnetic energy from one cycle to the next.

I_{MIN}>0

D_Z=0

The current ripple remains:

\Delta I_P=I_{MAX}-I_{MIN}=\frac{V_{P,ON}D}{L_m f_s}

The peak current can be written relative to the average current during the primary ON interval:

I_{MAX}=I_{AVG,ON}+\frac{\Delta I_P}{2}

I_{MIN}=I_{AVG,ON}-\frac{\Delta I_P}{2}

Incremental Energy Transfer in CCM

The transformer already contains energy at the beginning of the cycle. The energy added during the ON interval is therefore the difference between the maximum and minimum stored-energy states:

\Delta E=\frac{1}{2}L_m\left(I_{MAX}^{2}-I_{MIN}^{2}\right)

For an ideal fixed-frequency operating point:

P_{IN}\approx\frac{1}{2}L_m\left(I_{MAX}^{2}-I_{MIN}^{2}\right)f_s

CCM RMS Current

For a linear current ramp from I_MIN to I_MAX during duty cycle D, the primary RMS current over the complete switching period is:

I_{P,RMS}=\sqrt{\frac{D}{3}\left(I_{MIN}^{2}+I_{MIN}I_{MAX}+I_{MAX}^{2}\right)}

The secondary current follows the corresponding decreasing waveform during the OFF interval and remains above zero at the next cycle boundary.

Typical CCM Advantages and Tradeoffs

CCM Tendency	Engineering Consequence
Lower peak current for a given power	Can reduce switch, winding, rectifier, and capacitor current stress.
Nonzero minimum current	Creates magnetic bias and stored energy at every cycle boundary.
Smaller current ripple ratio	Can reduce RMS current and improve utilization of copper and semiconductor current capability.
Harder secondary-current commutation	Can increase diode reverse recovery or synchronous-rectifier timing difficulty.
Right-half-plane zero in the control-to-output response	Restricts achievable feedback-loop bandwidth and complicates compensation.
Greater stored energy under fault conditions	Makes current limiting, saturation margin, and protection timing especially important.

6. Comparing DCM, BCM, and CCM Waveforms

The most reliable way to identify operating mode is to align the gate-drive, physical primary-current, secondary-current, and primary-referred magnetizing-current waveforms over one switching period.

Waveform Feature	DCM	BCM	CCM
Primary-referred current at next turn-on	Zero	Approximately zero	Greater than zero
Secondary current reaches zero	Before next cycle	At next-cycle boundary	No
Zero-current interval	Present	Approximately zero	Absent
Peak current tendency	Highest for comparable power and frequency	Boundary value	Usually lower
Switching frequency	Fixed or variable	Often variable	Often fixed, but not required
Diode turn-off condition	Zero current	Zero current at boundary	Nonzero current commutation
Magnetic bias	No intentional residual current	No intentional residual current	Nonzero residual bias

These are general tendencies, not universal guarantees. Device technology, turns ratio, clamp topology, switching frequency, control algorithm, and operating point can change the practical ranking of efficiency, EMI, and component size.

Comparison of DCM, BCM, and CCM flyback operation showing primary current and core flux waveforms, including current reset, residual current, ripple, and flux swing.

7. Peak Current, RMS Current, and Copper Loss

Mode selection changes the current waveform used to size the primary conductor, secondary conductor, MOSFET, rectifier, current-sense network, clamp, and output capacitor. Average output power alone is not enough to establish those stresses.

DCM and BCM process energy from approximately zero current to a peak each cycle. CCM processes the required incremental energy around a nonzero current bias. This usually allows a lower peak current for the same power but introduces current at the beginning and end of every switching interval.

Primary Copper Loss

P_{CU,P}\approx I_{P,RMS}^{2}R_{P,AC}

Secondary Copper Loss

P_{CU,S}\approx I_{S,RMS}^{2}R_{S,AC}

The applicable AC resistance depends on frequency, harmonic content, conductor geometry, number of layers, proximity effect, winding arrangement, and gap fringing. A lower peak-current mode can still perform poorly if it requires more turns, a less favorable winding arrangement, or higher AC resistance.

SolidMag Engineering Insight

Current Waveform Matters More Than a Single Current Rating

A transformer current rating stated only as “5 A” is incomplete. The winding temperature depends on RMS current and AC resistance, saturation margin depends on the peak and bias current, and the clamp and semiconductor stresses depend on the switching transition.

Mode selection should therefore be evaluated with the complete current waveform—not one average or peak number taken out of context.

8. Flux Density, Core Bias, and Air-Gap Implications

The primary-referred current determines the magnetic operating point. In DCM and BCM, current begins near zero, so peak flux is commonly estimated from the peak current:

B_{PK}\approx\frac{L_m I_{P,PK}}{N_P A_e}

In CCM, the current ripple determines the AC flux excursion, while the nonzero minimum current establishes a magnetic bias:

\Delta B=\frac{L_m\left(I_{MAX}-I_{MIN}\right)}{N_P A_e}

B_{MIN}\approx\frac{L_m I_{MIN}}{N_P A_e}

B_{MAX}\approx\frac{L_m I_{MAX}}{N_P A_e}

A CCM design can have a smaller AC flux swing than a DCM design while still reaching a high absolute peak flux because of the DC bias. Core loss is driven strongly by the AC flux waveform, frequency, duty cycle, temperature, and material, while saturation margin depends on the absolute magnetic operating point.

The air gap, turns, magnetizing inductance, current limit, and core area must therefore be coordinated for the intended mode and for any mode transitions that occur across the operating range.

ENGINEERING CAUTION — DO NOT CHECK ONLY ΔB IN CCM

A CCM design can pass an AC flux-swing calculation and still approach saturation because the entire waveform is shifted by residual current. Verify the absolute maximum flux at worst-case current, minimum effective core area, maximum temperature, and minimum expected gap or inductance tolerance.

Also verify that transient current-limit behavior does not move the core into an unacceptable nonlinear region before protection responds.

9. Semiconductor, Capacitor, and Clamp Stress

Conduction mode affects much more than the transformer. It changes how current is commutated in the MOSFET, rectifier, output capacitor, input capacitor, and clamp network.

Design Area	DCM / BCM Consideration	CCM Consideration
Primary MOSFET current	Higher peak current and turn-off leakage energy can dominate stress.	Lower peak current is possible, but turn-on occurs with residual current and stored energy.
Secondary rectifier	Current naturally reaches zero, reducing reverse recovery.	Current commutates while nonzero, increasing diode recovery or synchronous-rectifier timing demands.
Output capacitor	Receives high pulsed current over a shorter conduction interval.	Often receives lower peak current over most or all of the OFF interval.
Input capacitor	Higher pulsed primary current can increase RMS current.	Lower peak current may reduce required RMS-current capability.
Clamp / snubber	Leakage energy scales with peak current squared.	Lower peak current can reduce leakage energy, but switching transitions remain important.
Current sensing	Requires adequate range for high DCM/BCM peaks.	Requires accurate sensing around a biased current waveform and reliable current limiting.

The leakage-inductance energy available at primary turn-off remains approximately:

E_{LK}=\frac{1}{2}L_{LK,P}I_{P,PK}^{2}

Because this energy depends on peak current squared, the higher peak current of a DCM or BCM design can materially increase clamp loss even when output power is unchanged.

10. Control Dynamics, Switching Frequency, and EMI

DCM Control Behavior

The usual fixed-frequency DCM control-to-output response behaves more like a first-order system because the transformer fully demagnetizes and each cycle begins from a known zero-current state. This can simplify compensation, although sampling effects, current-mode slope compensation, output-capacitor dynamics, and controller-specific behavior still require analysis.

CCM Control Behavior

CCM flyback operation introduces a right-half-plane zero in the control-to-output response. Increasing duty cycle initially stores more energy in the transformer but can delay delivery to the output, producing a nonminimum-phase response. The loop crossover frequency must remain sufficiently below that zero and other power-stage limits.

BCM and Variable Frequency

A BCM controller often waits for demagnetization and then turns the MOSFET on at a selected drain-voltage valley. This can reduce switching loss, but the frequency changes with line, load, inductance, and selected valley. The transformer must be checked over the full frequency range rather than at one nominal frequency.

EMI Is a Mode-and-Implementation Problem

DCM and BCM can reduce reverse-recovery noise but often create higher peak current and larger current slopes. CCM can reduce peak current but increases hard current commutation. Variable-frequency operation can distribute spectral energy while complicating filter design and compliance measurements. The final EMI result depends on transformer capacitance, leakage inductance, switch-node edge rate, clamp behavior, PCB layout, shielding, and filtering.

Infographic showing flyback operating mode tradeoffs and selection criteria for DCM, BCM, and CCM, including current behavior, flux swing, advantages, disadvantages, and application guidance.

11. Mode Transitions Across Line and Load

A real converter may not remain in one conduction mode. At light load, the current command decreases and a fixed-frequency design often enters DCM. At heavier load, current may reach the next cycle before fully demagnetizing, producing CCM. BCM lies at the transition between those conditions.

Input voltage also changes the current ramp, duty cycle, reflected-voltage timing, and demagnetization interval. A design can therefore be DCM at high line, CCM at low line, or vice versa depending on the controller and magnetizing inductance.

Multimode controllers may intentionally use:

CCM at high load to reduce peak current and conduction loss.
DCM or valley switching at moderate load to reduce switching loss.
Burst or skip operation at very light load to reduce standby power.
Frequency clamping or current limiting to keep magnetic and semiconductor stresses within bounds.

MODE TRANSITION IS NOT THE SAME AS BURST MODE

DCM means the magnetizing current reaches zero during an individual switching cycle. Burst or skip mode means the controller intentionally omits entire switching cycles or groups cycles.

A converter can operate in DCM during each active burst pulse while still using a separate burst-mode envelope at light load.

12. Worked Comparison — Boundary and CCM Designs at the Same Power

This idealized example compares two magnetic designs for the same 48 V input, 12 V / 2 A output, 100 kHz operating point. It also shows how the boundary design enters DCM at approximately half load. The calculations neglect losses except for an assumed 90% efficiency and use a 4:1 primary-to-secondary turns ratio with a 0.5 V secondary rectifier drop.

WORKED-EXAMPLE ASSUMPTIONS

Input voltage: 48 V. Output power: 24 W. Estimated efficiency: 90%. Switching frequency at the comparison point: 100 kHz. Turns ratio: 4:1. Reflected voltage: 50 V. Idealized linear current ramps and constant voltages are assumed.

The example compares current and timing relationships only. It does not complete core selection, gap design, winding loss, clamp design, control compensation, thermal analysis, or semiconductor derating.

Step 1 — Calculate Input Power and Duty Cycle

P_{IN}=\frac{24}{0.90}=26.67\ \mathrm{W}

V_R=4\left(12+0.5\right)=50\ \mathrm{V}

D=\frac{V_R}{V_{IN}+V_R}=\frac{50}{48+50}=0.510

D_S=1-D=0.490

Step 2 — Boundary-Mode Magnetizing Inductance

L_{m,BCM}=\frac{48^{2}\left(0.510\right)^{2}}{2\left(26.67\right)\left(100000\right)}=112.5\ \mu\mathrm{H}

I_{P,PK}=\frac{48\left(0.510\right)}{\left(112.5\ \mu\mathrm{H}\right)\left(100000\right)}=2.18\ \mathrm{A}

I_{P,RMS}=2.18\sqrt{\frac{0.510}{3}}=0.90\ \mathrm{A}

At this operating point the current reaches zero at the end of the secondary conduction interval, so the design is at the DCM/CCM boundary.

Step 3 — CCM Alternative with Twice the Magnetizing Inductance

Now consider a CCM magnetic design with approximately twice the magnetizing inductance while delivering the same input power at the same voltage, duty cycle, and frequency:

L_{m,CCM}=224.9\ \mu\mathrm{H}

\Delta I_P=\frac{48\left(0.510\right)}{\left(224.9\ \mu\mathrm{H}\right)\left(100000\right)}=1.09\ \mathrm{A}

I_{AVG,ON}=\frac{P_{IN}}{V_{IN}D}=1.09\ \mathrm{A}

I_{MIN}=1.09-\frac{1.09}{2}=0.54\ \mathrm{A}

I_{MAX}=1.09+\frac{1.09}{2}=1.63\ \mathrm{A}

I_{P,RMS}=0.81\ \mathrm{A}

The CCM alternative reduces primary peak and RMS current at this operating point, but it carries residual magnetic bias and requires CCM control, current limiting, rectifier commutation, and loop compensation to be designed accordingly.

Step 4 — Half-Load DCM Operation of the Boundary Design

Using the 112.5 µH boundary transformer at approximately half input power, the required duty cycle decreases to about 0.361 at 100 kHz:

D_{DCM}\approx0.361

I_{P,PK}\approx1.54\ \mathrm{A}

D_S\approx0.346

D_Z=1-D-D_S\approx0.293

The same transformer that operates at the boundary near full load therefore operates in DCM at this lighter-load point, with nearly 29% of the switching period remaining as zero-current time.

Quantity	Half-Load DCM, 112.5 µH	Full-Load BCM, 112.5 µH	Full-Load CCM, 224.9 µH
Input power	13.33 W	26.67 W	26.67 W
Duty cycle	0.361	0.510	0.510
Minimum primary-referred current	0 A	≈ 0 A	0.54 A
Peak primary current	1.54 A	2.18 A	1.63 A
Primary RMS current	0.53 A	0.90 A	0.81 A
Zero-current fraction	0.293	≈ 0	0
Operating mode	DCM	BCM	CCM

SolidMag Engineering Insight

One Transformer Can Cross Multiple Modes

Conduction mode is not fixed by the transformer name. The same magnetizing inductance can operate in DCM at light load, pass through BCM, and enter CCM as power increases or operating conditions change.

The transformer, current limit, rectifier, clamp, control loop, and thermal design must be valid across every mode the product is allowed to enter.

13. Selecting the Appropriate Operating Mode

The best conduction mode depends on the complete converter rather than on one universal power threshold. Consider the following questions:

What are the minimum, nominal, and maximum input voltages?
What power and transient load must be delivered?
Is fixed frequency required, or is variable-frequency valley switching acceptable?
How much peak current can the MOSFET, transformer, rectifier, clamp, and current-sense path tolerate?
How important are diode reverse recovery and synchronous-rectifier timing?
What feedback bandwidth and transient response are required?
What transformer size, copper loss, core loss, EMI, audible-noise, and cost limits apply?
Will the controller transition between modes across line and load?

Design Priority	Mode Often Considered First	Reason for Evaluation
Low-power, simple, low-cost converter	DCM	Simple dynamics and zero-current secondary turn-off can be attractive.
Medium-power design with reduced peak current	CCM	Lower peak and RMS current may improve semiconductor and copper utilization.
Valley switching and zero-current demagnetization	BCM / CrM / QR-DCM	Variable-frequency control can reduce turn-on loss and diode recovery.
Wide load range and high efficiency	Multimode	The controller can use different modes where each is most beneficial.
Tight loop bandwidth requirement	DCM or carefully compensated CCM	CCM right-half-plane-zero behavior can restrict bandwidth.

Chapter Summary

DCM has a finite zero-current interval before the next cycle.
BCM occurs when current reaches zero at the next switching boundary.
CCM carries nonzero residual magnetizing current into the next cycle.
DCM and BCM generally use higher peak current; CCM generally reduces peak current but adds magnetic bias and harder current commutation.
Operating mode changes flux bias, RMS current, copper loss, semiconductor stress, control dynamics, EMI, and transformer design.
A converter may transition between modes across line and load, so all permitted modes must be verified.

SolidMag Engineering Insight

Choose the Mode That Optimizes the Complete Converter

DCM, BCM, and CCM are not quality rankings. Each mode redistributes stress among magnetics, semiconductors, capacitors, control dynamics, switching loss, and EMI.

The best choice is the mode—or controlled combination of modes—that satisfies efficiency, size, cost, thermal, transient, safety, and manufacturability requirements over the complete operating envelope.

Technical References for This Chapter

Related SolidMagnetics Resources

Chapter 2 — Energy Transfer

Review How Flyback Energy Transfer Works for the ON/OFF current paths, winding polarity, reflected voltage, and switching-cycle waveforms.

Complete Flyback Transformer Design Guide

Use the flagship guide for turns, cores, air gaps, conductors, insulation, leakage, losses, thermal design, and the complete worked example.

Chapter 4 — Design Requirements

Continue to defining flyback transformer design requirements, including line, load, isolation, thermal, mechanical, compliance, and optimization inputs.

Flyback Transformer Designer

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DESIGN ASSUMPTION — IDEALIZED MODE COMPARISON

1. What Conduction Mode Means in a Flyback Converter

Operating Mode Is a Property of the Current Waveform

2. The Magnetizing-Current Waveform Defines the Mode

3. Discontinuous Conduction Mode — DCM

DCM Peak Current and Energy

DCM RMS Currents

Typical DCM Advantages and Tradeoffs

ENGINEERING CAUTION — DCM DOES NOT AUTOMATICALLY MEAN LOW EMI

4. Boundary Conduction Mode — BCM, CrM, or Transition Mode

Boundary Magnetizing Inductance at a Defined Operating Point

Typical BCM Advantages and Tradeoffs

BCM Is a Boundary Condition, Not a Wide Operating Region

5. Continuous Conduction Mode — CCM

Incremental Energy Transfer in CCM

CCM RMS Current

Typical CCM Advantages and Tradeoffs

6. Comparing DCM, BCM, and CCM Waveforms

7. Peak Current, RMS Current, and Copper Loss

Primary Copper Loss

Secondary Copper Loss

Current Waveform Matters More Than a Single Current Rating

8. Flux Density, Core Bias, and Air-Gap Implications

ENGINEERING CAUTION — DO NOT CHECK ONLY ΔB IN CCM

9. Semiconductor, Capacitor, and Clamp Stress

10. Control Dynamics, Switching Frequency, and EMI

DCM Control Behavior

CCM Control Behavior

BCM and Variable Frequency

EMI Is a Mode-and-Implementation Problem

11. Mode Transitions Across Line and Load

MODE TRANSITION IS NOT THE SAME AS BURST MODE

12. Worked Comparison — Boundary and CCM Designs at the Same Power

WORKED-EXAMPLE ASSUMPTIONS

Step 1 — Calculate Input Power and Duty Cycle

Step 2 — Boundary-Mode Magnetizing Inductance

Step 3 — CCM Alternative with Twice the Magnetizing Inductance

Step 4 — Half-Load DCM Operation of the Boundary Design

One Transformer Can Cross Multiple Modes

13. Selecting the Appropriate Operating Mode

Chapter Summary

Choose the Mode That Optimizes the Complete Converter

Technical References for This Chapter

Related SolidMagnetics Resources

Chapter 2 — Energy Transfer

Complete Flyback Transformer Design Guide

Chapter 4 — Design Requirements

Flyback Transformer Designer

Ready to Design a Flyback Transformer?

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Stan Gibson